Wireless terminal

ABSTRACT

A wireless terminal a transceiver coupled to an antenna feed and a ground conductor ( 502 ), the antenna feed being coupled directly to the ground conductor ( 502 ). In one embodiment the ground conductor is a conducting case ( 902 ). The coupling is via a parallel plate capacitor formed by a respective plate ( 506 ) and a portion of the surface of the case ( 502 ). The case ( 502 ) acts as an efficient, wideband radiator, eliminating the need for separate antennas. Slots ( 912, 1214 ) perform a matching function, eliminating the need for matching between the transceiver and antenna feed.

[0001] The present invention relates to a wireless terminal, for examplea mobile phone handset.

[0002] Wireless terminals, such as mobile phone handsets, typicallyincorporate either an external antenna, such as a normal mode helix ormeander line antenna, or an internal antenna, such as a PlanarInverted-F Antenna (PIFA) or similar.

[0003] Such antennas are small (relative to a wavelength) and therefore,owing to the fundamental limits of small antennas, narrowband. However,cellular radio communication systems typically have a fractionalbandwidth of 10% or more. To achieve such a bandwidth from a PIFA forexample requires a considerable volume, there being a directrelationship between the bandwidth of a patch antenna and its volume,but such a volume is not readily available with the current trendstowards small handsets. Hence, because of the limits referred to above,it is not feasible to achieve efficient wideband radiation from smallantennas in present-day wireless terminals.

[0004] A further problem with known antenna arrangements for wirelessterminals is that they are generally unbalanced, and therefore couplestrongly to the terminal case. As a result a significant amount ofradiation emanates from the terminal itself rather than the antenna. Awireless terminal in which an antenna feed is directly coupled to theterminal case, thereby taking advantage of this situation, is disclosedin our co-pending unpublished International patent applicationPCT/EPO1/08550 (Applicant's reference PHGB010056). When fed via anappropriate matching network the terminal case acts as an efficient,wideband radiator.

[0005] An object of the present invention is to provide a compactwireless terminal having efficient radiation properties without the needfor a matching network.

[0006] According to the present invention there is provided a wirelessterminal comprising a ground conductor and a transceiver coupled to anantenna feed, wherein the antenna feed is coupled directly to the groundconductor via a capacitor formed by a conducting plate and a portion ofthe ground conductor and wherein a slot, partially located underneaththe conducting plate, is provided in the ground conductor.

[0007] The location of a slot beneath the conducting plate performs muchof the function of a conventional matching circuit, thereby simplifyingimplementation of a wireless terminal. More than one slot may beprovided, and a slot may be folded as dictated by space or otherrequirements.

[0008] The present invention is applicable to any wireless communicationsystem where the use of a large antenna is not appropriate. Since thecoupling capacitor is small, it is ideally suited to an RF IC or module,where the coupling capacitor would be part of the module. It isparticularly useful in wireless systems that feature multiband orwideband operation.

[0009] The present invention is based upon the recognition, not presentin the prior art, that the impedances of an antenna and a wirelesshandset are similar to those of an asymmetric dipole, which areseparable, and on the further recognition that the antenna impedance canbe replaced with a non-radiating coupling element.

[0010] Embodiments of the present invention will now be described, byway of example, with reference to the accompanying drawings, wherein:

[0011]FIG. 1 shows a model of an asymmetrical dipole antenna,representing the combination of an antenna and a wireless terminal;

[0012]FIG. 2 is a graph demonstrating the separability of the componentsof the impedance of an asymmetrical dipole;

[0013]FIG. 3 is an equivalent circuit of the combination of a handsetand an antenna;

[0014]FIG. 4 is an equivalent circuit of a capacitively back-coupledhandset;

[0015]FIG. 5 is a perspective view of a basic capacitively back-coupledhandset;

[0016]FIG. 6 is a graph of simulated return loss S₁₁ in dB againstfrequency f in MHz for the handset of FIG. 5;

[0017]FIG. 7 is a Smith chart showing the simulated impedance of thehandset of FIG. 5 over the frequency range 1000 to 2800 MHz;

[0018]FIG. 8 is a graph showing the simulated resistance of the handsetof FIG. 5;

[0019]FIG. 9 is a plan view of a single-slotted self-resonantcapacitively back-coupled handset;

[0020]FIG. 10 is a graph of simulated return loss S₁₁ in dB againstfrequency f in MHz for the handset of FIG. 9;

[0021]FIG. 11 is a Smith chart showing the simulated impedance of thehandset of FIG. 9 over the frequency range 800 to 3000 MHz;

[0022]FIG. 12 is a plan view of a doubly-slotted self-resonantcapacitively back-coupled handset;

[0023]FIG. 13 is a graph of simulated return loss S₁₁ in dB againstfrequency f in MHz for the handset of FIG. 12;

[0024]FIG. 14 is a Smith chart showing the simulated impedance of thehandset of FIG. 12, over the frequency range 800 to 3000 MHz;

[0025]FIG. 15 is a graph of simulated return loss S₁₁ in dB againstfrequency f in MHz for the handset of FIG. 12 fed via a matchingnetwork; and

[0026]FIG. 16 is a Smith chart showing the simulated impedance of thehandset of FIG. 12 fed via a matching network, over the frequency range800 to 3000 MHz.

[0027] In the drawings the same reference numerals have been used toindicate corresponding features.

[0028]FIG. 1 shows a model of the impedance seen by a transceiver, intransmit mode, in a wireless handset at its antenna feed point. Theimpedance is modelled as an asymmetrical dipole, where the first arm 102represents the impedance of the antenna and the second arm 104 theimpedance of the handset, both arms being driven by a source 106. Asshown in the figure, the impedance of such an arrangement issubstantially equivalent to the sum of the impedance of each arm 102,104driven separately against a virtual ground 108. The model could equallywell be used for reception by replacing the source 106 by an impedancerepresenting that of the transceiver, although this is rather moredifficult to simulate.

[0029] The validity of this model was checked by simulations using thewell-known NEC (Numerical Electromagnetics Code) with the first arm 102having a length of 40 mm and a diameter of 1 mm and the second arm 104having a length of 80 mm and a diameter of 1 mm. FIG. 2 shows theresults for the real and imaginary parts of the impedance (R+jX) of thecombined arrangement (Ref R and Ref X) together with results obtained bysimulating the impedances separately and summing the result. It can beseen that the results of the simulations are quite close. The onlysignificant deviation is in the region of half-wave resonance, when theimpedance is difficult to simulate accurately.

[0030] An equivalent circuit for the combination of an antenna and ahandset, as seen from the antenna feed point, is shown in FIG. 3. R₁ andjX₁ represent the impedance of the antenna, while R₂ and jX₂ representthe impedance of the handset. From this equivalent circuit it can bededuced that the ratio of power radiated by the antenna, P₁, and thehandset, P₂, is given by $\frac{P_{1}}{P_{2}} = \frac{R_{1}}{R_{2}}$

[0031] If the size of the antenna is reduced, its radiation resistanceR₁ will also reduce. If the antenna becomes infinitesimally small itsradiation resistance R₁ will fall to zero and all of the radiation willcome from the handset. This situation can be made beneficial if thehandset impedance is suitable for the source 106 driving it and if thecapacitive reactance of the infinitesimal antenna can be minimised byincreasing the capacitive back-coupling to the handset.

[0032] With these modifications, the equivalent circuit is modified tothat shown in FIG. 4. The antenna has therefore been replaced with aphysically very small back-coupling capacitor, designed to have a largecapacitance for maximum coupling and minimum reactance. The residualreactance of the back-coupling capacitor can be tuned out with a simplematching circuit. By correct design of the handset, the resultingbandwidth can be much greater than with a conventional antenna andhandset combination, because the handset acts as a low Q radiatingelement (simulations show that a typical Q is around 1), whereasconventional antennas typically have a Q of around 50.

[0033] A basic embodiment of a capacitively back-coupled handset isshown in FIG. 5. A handset 502 has dimensions of 10×40×100 mm, typicalof modern cellular handsets. A parallel plate capacitor 504, havingdimensions 2×10×10 mm, is formed by mounting a 10×10 mm plate 506 2 mmabove the top edge 508 of the handset 502, in the position normallyoccupied by a much larger antenna. The resultant capacitance is about0.5 pF, representing a compromise between capacitance (which would beincreased by reducing the separation of the handset 502 and plate 506)and coupling effectiveness (which depends on the separation of thehandset 502 and plate 506). The capacitor is fed via a support 510,which is insulated from the handset case 502.

[0034] The return loss S₁₁ of this embodiment after matching wassimulated using the High Frequency Structure Simulator (HFSS), availablefrom Ansoft Corporation, with the results shown in FIG. 6 forfrequencies f between 1000 and 2800 MHz. A conventional two inductor “L”network was used to match at 1900 MHz. The resultant bandwidth at 7 dBreturn loss (corresponding to approximately 90% of input power radiated)is approximately 60 MHz, or 3%, which is useful but not as large as wasrequired. A Smith chart illustrating the simulated impedance of thisembodiment over the same frequency range is shown in FIG. 7.

[0035] The low bandwidth is because the combination of the handset 502and capacitor 504 present an impedance of approximately 3−j90 Ω at 1900MHz. FIG. 8 shows the resistance variation, over the same frequencyrange as before, simulated using HFSS. This can be improved byredesigning the case to increase the resistance, for example by the useof a slot or a narrower handset, as discussed in our co-pendingunpublished International patent application PCT/EP01/08550.

[0036] The handset of FIG. 5 requires matching to obtain reasonableperformance. There are significant advantages to being able to eliminatethe need for matching. A plan view of a modified single bandconfiguration which requires no matching is shown in FIG. 9. Thisembodiment differs from that of FIG. 5 in that the 10 mm square plate506 is located 2 mm above the back of the handset 502, and in that aslot 912 of length 30 mm and width 1 mm is cut in the conductingmaterial 2 mm from the edge of the handset case. The slot 912 extendsunder the conducting plate 506 (as shown by dashed lines in FIG. 9). Theslot 912 is resonant at odd multiples of a quarter wavelength, i.e. atλ/4, 3 λ/4, etc.

[0037] The slot presents a high impedance to the coupling capacitor,thereby enabling a good match to 50 Ω. It is believed that the capacitorexcites a transmission line mode in the slot 912 that acts as a shuntinductance at the antenna feed, which acts to match the response.

[0038] In the illustrated embodiment the slot 912 is located close tothe edge of the handset case 502 in order to minimise the space used,although the slot could equally well be located on the other side of thecoupling capacitor 504. Similarly, the coupling capacitor could beimplemented in other positions on the handset 502 and the slot 912 couldhave a range of configurations, for example vertical, horizontal ormeandering.

[0039] The return loss S₁₁ of this embodiment, without matching, wassimulated using HFSS, with the results shown in FIG. 10 for frequenciesf between 800 and 3000 MHz. The resultant bandwidth at 7 dB return lossis approximately 90 MHz, or 4.3%. Although the bandwidth could beimproved with matching, it is useful to be able to avoid having toinclude matching and the bandwidth is already more than sufficient for aBluetooth embodiment, for example.

[0040] A Smith chart illustrating the simulated impedance of thisembodiment over the same frequency range is shown in FIG. 11. This showsthat the configuration of FIG. 9 also has the useful property thatresonance (zero reactance) is achieved twice, with the higher frequencyresonance having the higher resistance. This is particularly convenient,since the receive band is usually at a higher frequency in a frequencyduplex system.

[0041] A preferred transceiver architecture is to maintain a lowimpedance path between the (generally low impedance) transmitter and theantenna, and a high impedance path between the antenna and the(generally high impedance) receiver. However, for simplicity of designit is conventional to use a 50 Ω system impedance with additionalmatching at the transmitter and receiver as required. This matching islossy, and may also reduce the bandwidth seen at both the transmitterand receiver. Hence, the removal of the need for matching is asignificant advantage of the present invention.

[0042] A dual band embodiment of the present invention is shown in planview in FIG. 12. In this embodiment the plate 506 and slot 912 have beenmoved to the top centre of the back surface of the handset 502, and afurther slot 1214 has been added. The further slot 1214 is longer thanthe first slot 912, having a a total length of approximately 73 mm and awidth of 1 mm, and folded to reduce the area it occupies.

[0043] The return loss S₁₁ of this embodiment, without matching, wassimulated using HFSS, with the results shown in FIG. 13 for frequenciesf between 800 and 3000 MHz. It can clearly be seen that this designallows dual, tri or multiband operation. The slots 912, 1214 areresonant at odd multiples of λ/4, and can therefore be arranged to giveindividual or combined resonances.

[0044] The first resonance (at approximately 1 GHz) is the λ/4 resonanceof the longer slot 1214. The second resonance (at approximately 1.8 GHz)is the λ/4 resonance of the shorter slot 912. The third resonance (atapproximately 2.8 GHz) is the 3 λ/4 resonance of the longer slot 1214.It is clear, for example, that, with some modification, thisconfiguration can be used for GSM, DCS1800 and Bluetooth.

[0045] The resultant bandwidths at 7 dB return loss for the threeresonances are approximately 15 MHz (1.5%), 110 MHz (5.9%) and 110 MHz(3.9%). The bandwidth of the 1 GHz resonance is small, but the otherbandwidths are good. A Smith chart illustrating the simulated impedanceof this embodiment over the same frequency range is shown in FIG. 13.The rapid changes in impedance in the Smith chart reflect thenarrow-band nature of the first resonance.

[0046] The self-resonance of each slot 912,1214 is independentlyvariable via its position under the feeding capacitor 504: as the slot912,1214 is progressively moved under the plate 506 the effect of itsnominal shunt inductance increases. Also, each slot 912,1214 is highimpedance at its open end and low impedance at its shorted end. Hence,the resistance could be varied by tapping off at various points alongthe slot. The capacitor can also be made asymmetric to allow for suchtapping to be performed, to some extent.

[0047] Embodiments of the present invention may also be used inconjunction with matching. As an example, simulations of the dual slotconfiguration illustrated in FIG. 12 in conjunction with a simple “L”matching circuit similar to that used for the basic embodiment of FIG. 5were performed. Results for the return loss S₁₁ are shown in FIG. 15 forfrequencies f between 800 and 3000 MHz. It can be seen that a very widebandwidth is achieved (a 3 dB bandwidth of approximately 1.4 GHz). Thiscould be enhanced further with a more elaborate matching circuit. ASmith chart illustrating the simulated impedance of this embodiment overthe same frequency range is shown in FIG. 16.

[0048] In the above embodiments a conducting handset case has been theradiating element. However, other ground conductors in a wirelessterminal could perform a similar function. Examples include conductorsused for EMC shielding and an area of Printed Circuit Board (PCB)metallisation, for example a ground plane.

[0049] From reading the present disclosure, other modifications will beapparent to persons skilled in the art. Such modifications may involveother features which are already known in the design, manufacture anduse of wireless terminals and component parts thereof, and which may beused instead of or in addition to features already described herein.

1. A wireless terminal comprising a ground conductor and a transceivercoupled to an antenna feed, wherein the antenna feed is coupled directlyto the ground conductor via a capacitor formed by a conducting plate anda portion of the ground conductor and wherein a slot, partially locatedunderneath the conducting plate, is provided in the ground conductor. 2.A terminal as claimed in claim 1, characterised in that the slot isparallel to the major axis of the terminal.
 3. A terminal as claimed inclaim 1, characterised in that the slot is folded.
 4. A terminal asclaimed in claim 1, characterised in that a further slot, also partiallylocated underneath the conducting plate, is provided in the groundconductor.
 5. A terminal as claimed in claim 1, characterised in thatthe conducting plate is asymmetrical with respect to the major axis ofthe ground conductor.
 6. A terminal as claimed in claim 1, characterisedin that the ground conductor is a handset case.
 7. A terminal as claimedin claim 1, characterised in that the ground conductor is a printedcircuit board ground plane.
 8. A terminal as claimed in claim 1,characterised in that a matching network is provided between thetransceiver and the antenna feed.